1. Field of the Invention
The present invention relates to a detection circuit using a differential capacitive sensor with input-common-mode control in a sense interface, in particular a sense interface of the fully differential switched-capacitor type, to which the following description will make reference, without this implying any loss of generality.
2. Description of the Related Art
The last few years have witnessed a widespread use of detection circuits that employ differential capacitive sensors (for example, inertial sensors, accelerometers, pressure or force sensors) in applications that envisage low supply voltages and low power consumption, such as for example in battery-supplied portable devices (PDAs, digital audio players, cell phones, digital camcorders and the like). As is known, capacitive differential sensors base their operation on a capacitive unbalancing, which occurs as a function of a quantity to be detected (an acceleration, a pressure, a force, etc.). In particular, there is a widespread use of micro-electromechanical-system (MEMS) sensors, obtained with techniques of microfabrication of semiconductor materials. In a known way, these sensors comprise a fixed body (“stator”) and a mobile mass (designated by the term “rotor”), both of which are generally made of appropriately doped semiconductor material and which are connected to one another by means of elastic elements (springs) and constrained so that the rotor has, with respect to the stator, pre-set translational and/or rotational degrees of freedom. The stator has a plurality of fixed arms, and the rotor has a plurality of mobile arms, said arms facing one another so as to form pairs of capacitors having a capacitance that varies as a function of the relative position of the arms, i.e., as a function of the relative position of the rotor with respect to the stator. Accordingly, when the sensor is affected by the quantity to be determined, the rotor shifts and a capacitive unbalancing of the pairs of capacitors occurs, from which it is possible to determine the desired quantity. According to the type of structure and the type of relative movement between rotor and stator, it is possible to provide MEMS sensors of a linear or rotational type, with variation of the gap (i.e., the distance between the mobile arms and the respective fixed arms) and/or with variation of a degree of facing (i.e., variation of the area of mutual facing between the mobile arms and the respective fixed arms).
Purely by way of example, FIG. 1a schematically illustrates a differential capacitive sensor 1, of a linear MEMS type. The following exposition is in any case to be understood as valid for MEMS sensors having different configurations. In detail, the differential capacitive sensor 1 comprises a stator, of which just the first fixed arms 2a and second fixed arms 2b are illustrated, and a rotor, constituted by a mobile mass 3 and by mobile arms 4 fixed to the mobile mass 3. Each mobile arm 4 is set between a respective first fixed arm 2a and a second fixed arm 2b. The mobile mass 3 is suspended via springs 5 to anchoring elements 6, and is mobile along an axis x that constitutes the preferential axis of detection of the differential capacitive sensor 1. The first fixed arms 2a and the second fixed arms 2b are electrically connected to a first stator terminal 7a and to a second stator terminal 7b, respectively, whilst the mobile arms 4 are electrically connected to a rotor terminal 8.
As illustrated in FIG. 1b, the differential capacitive sensor 1 has an equivalent electrical circuit comprising a first sense capacitor 9a and a second sense capacitor 9b, with plane and parallel faces, arranged in “half-bridge” configuration, i.e., connected in series between the first stator terminal 7a and the second stator terminal 7b, and having in common the rotor terminal 8. The capacitances of the first sense capacitor 9a and of the second sense capacitor 9b are variable as a function of the distance between the mobile arms 4 and the fixed arms 2a, 2b, and thus as a function of the displacement of the rotor with respect to the stator. In particular, the first sense capacitor 9a is the parallel of the capacitances formed between the first fixed arms 2a and the mobile arms 4, whilst the second sense capacitor 9b is the parallel of the capacitances formed between the second fixed arms 2b and the mobile arms 4. When the differential capacitive sensor 1 is subjected to an acceleration along the axis x, the mobile mass 3 moves along this axis, and consequently a capacitive variation of the first sense capacitor 9a and a capacitive variation of the second sense capacitor 9b are produced, said variations being equal in absolute value and opposite in sign with respect to one another. In particular, given a common sense capacitance Cs at rest for the first sense capacitor 9a and the second sense capacitor 9b (assuming the differential capacitive sensor 1 as being symmetrical at rest), due to unbalancing, the first sense capacitor 9a assumes a value of capacitance equal to Cs1=Cs+ΔCs, and the second sense capacitor 9b assumes a value of capacitance equal to Cs2=Cs−ΔCs.
As is known, in the aforesaid detection circuits, an appropriate sense circuit is coupled to the differential capacitive sensor, and usually comprises a charge-integrator interface stage (or charge-amplifier, operating as charge-to-voltage converter), and appropriate stages of amplification, filtering and noise canceling, cascaded to the interface stage. The sense circuit applies a read pulse (having a voltage in the region of a few volts) to the rotor terminal, reads the resultant capacitive unbalancing ΔCs, and generates, from said capacitive unbalancing, an output electrical signal correlated to the quantity that is to be detected. In applications with low supply voltage and low consumption, the performance required to the sense circuit in terms of resolution and of thermal and long-term (aging) stability are particularly stringent, and call for the development of read techniques that are as immune as possible from error, such as noise (thermal noise and low-frequency noise) and offset. For this reason, recently-see for example the article “A Three-Axis Micromachined Accelerometer with a CMOS Position-Sense Interface and Digital Offset-Trim Electronics” by M. Lemkin, B. E. Boser, IEEE Journal of Solid-State Circuits, Vol. 34, No. 4, April 1999, pp. 456-468 which is considered included herein in its entirety—the use of fully differential sense circuits of the switched-capacitor type (operating in discrete-time) has been proposed, which make it possible to operate at low supply voltages and which intrinsically meet the need of current consumption reduction. In particular, the use in the detection circuit of differential sensors coupled to fully differential sense circuits enables numerous advantages to be obtained, amongst which: the increase in the rejection of noise coming from the supply (and/or from the substrate in case of integrated technologies); the reduction of errors, such as charge injection or the so-called “dock feedthrough” (the latter being intrinsically due to the use of switches); and the increase in the dynamics of the output signals by a factor of two. However, a problem linked to the use of fully differential circuits regards the need to eliminate or at least limit the effects due to the common-mode signal at their input. In particular, the read pulse applied to the rotor terminal (which is chosen as wide as possible compatibly with the design requirements, for the purpose of increasing the signal-to-noise ratio at the output of the sense stage) produces a common-mode signal at the inputs of the charge integrator of the interface stage. Said common-mode signal is caused by a common-mode amount of charge (i.e., an amount of charge that is the same on both of the inputs of the charge integrator) injected by the first sense capacitor 9a and by the second sense capacitor 9b following application of the read pulse. The common-mode signal must be cancelled in order to reduce the read errors that can derive therefrom, in particular a gain error and offset error depending on a mismatch of parasitic capacitances (in particular, the “pad” capacitance and substrate capacitance) at the input terminals of the charge integrator. To solve this problem, in the aforesaid article it is proposed to implement an input-common-mode control circuit of an active type, which uses a feedback loop (so-called ICMFB-Input-Common-Mode Feedback).
The above solution is now described briefly with reference to FIG. 2, which shows a detection circuit comprising an interface circuit 10, of the fully differential switched-capacitor type, coupled to the differential capacitive sensor 1, represented schematically, in accordance with what has been described previously, with the first sense capacitor 9a and the second sense capacitor 9b having sense capacitance at rest Cs, and having first terminals connected together and to the rotor terminal 8, and second terminals connected, respectively, to the first stator terminal 7a and to the second stator terminal 7b. The interface circuit 10 is connected at input to the first stator terminal 7a and to the second stator terminal 7b and comprises a charge integrator 12 and a feedback stage 14 implementing the ICMFB active circuit for input-common-mode control. The parasitic capacitances are represented schematically as a first parasitic capacitor 15 and a second parasitic capacitor 16, connected, respectively, between the first stator terminal 7a and the second stator terminal 7b and a reference-potential line 18 (coinciding, in particular, with the signal ground), and having parasitic capacitance Cp.
In detail, the charge integrator 12 comprises a sense operational amplifier 20, in charge-integrator configuration (which carries out a conversion of an input charge into an output voltage), which has an inverting input connected to the first stator terminal 7a and a non-inverting input connected to the second stator terminal 7b, and two outputs, between which an output voltage Vo is present. The charge integrator 12 further comprises a first integration capacitor 22 and a second integration capacitor 23, having the same integration capacitance Ci and connected the first between the inverting input and an output, and the second between the non-inverting input and the other output of the sense operational amplifier 20.
The feedback stage 14 comprises an amplifier circuit 25, and a first feedback capacitor 26 and a second feedback capacitor 27 having the same feedback capacitance Cfb. The amplifier circuit 25, the structure and operation of which are described in detail in the article referred to above, is a switched-capacitor circuit having an output 25a, a first differential input 25b and a second differential input 25c, which are connected, respectively, to the inverting input and to the non-inverting input of the sense operational amplifier 20, and a reference input 25d, connected to the reference-potential line 18. The first feedback capacitor 26 and the second feedback capacitor 27 have first terminals connected to one another and to the output 25a of the amplifier circuit 25, and second terminals connected to the first stator terminal 7a and to the second stator terminal 7b, respectively. In use, the amplifier circuit 25 detects the voltage between the first differential input 25b and the second differential input 25c, determines its mean value, and generates at the output 25a a feedback voltage Vfb proportional to the difference between said mean value and the reference voltage of the reference-potential line 18.
Reading of the differential capacitive sensor 1 is obtained by supplying to the rotor terminal 8 (and to the mobile mass 3) a step read signal Vr (which has a voltage variation having a value, for example, equal to the supply voltage of the interface circuit 10, or else equal to a fraction of said supply voltage). The charge integrator 12 integrates the differential amount of charge supplied by the first sense capacitor 9a and by the second sense capacitor 9b (i.e., caused by the capacitive unbalancing ΔCs of the two capacitors), and consequently generates the output voltage Vo. In particular, the following relation of proportionality is valid for the output voltage Vo:
      V    o    ⁢      αV    r    ⁢            Δ      ⁢                          ⁢              C        s                    C      i      where, as mentioned previously, ΔCs is the capacitive unbalancing of the differential capacitive sensor 1, i.e., the equal and opposite variation of capacitance of the first sense capacitor 9a and of the second sense capacitor 9b, which occurs due to displacements of the mobile mass 4 with respect to the stator. The feedback stage 14, through the feedback voltage Vfb, keeps the first stator terminal 7a and the second stator terminal 7b at a constant common-mode voltage with respect to the reference voltage. Furthermore, since the sense operational amplifier 20 keeps the voltage between its inputs substantially at zero, the first stator terminal 7a and the second stator terminal 7b are practically virtual-ground points. In this way, the influence of the parasitic capacitors 15, 16 on the sense circuit is eliminated, in so far as they are kept at a constant voltage and consequently do not absorb electric charge.
However, even though the feedback stage 14 is advantageous in so far as it enables elimination of the common-mode problems, it should be designed taking into account the requirements of low supply voltage and of low power consumption. In particular, the output dynamics of the amplifier circuit 25 should be lower than the supply voltage of the interface circuit 10; for example, it may be equal to one third of said supply voltage. It follows that the feedback capacitance Cfb of the feedback capacitors 26, 27 should be greater than the sense capacitance at rest Cs of the differential capacitive sensor 1; for example, it may be equal to three times said value, in the case where the voltage variation of the read signal Vr is equal to the supply voltage. As described in the article referred to, however, the fluctuations of voltage at the input of the sense operational amplifier 20 due to the noise cause a flow of charge in the integration capacitors 22, 23, which comes both from the sense capacitors 9a, 9b and from the parasitic capacitors 15, 16, causing the noise to be amplified by a factor equal to:
                    V        o        2            _                      V                  o          ⁢                                          ⁢          p                2            _        =            (              1        +                                            C              s                        +                          C              p                        +                          C                              f                ⁢                                                                  ⁢                b                                                          C            i                              )        2  where Vop is the value of an equivalent input-noise generator. Consequently, it is evident from said relation that the noise at output from the charge integrator 12 increases quadratically as the value of the feedback capacitance Cfb increases. Consequently, said value should be as contained as possible in order to reduce the output noise (or equivalently the current consumption given the same noise). The introduction of the ICMFB circuit, which itself involves a non-negligible current consumption (on account of the presence of amplifier components), although solving the problem linked to the common mode, because of the high value of the feedback capacitance Cfb risks worsening the noise performance (or further increasing the current consumption, given the same noise) of the sense interface. The current consumption may even be excessive for portable applications.